Apparatus for transmitting broadcast signals, apparatus for receiving broadcast signals, method for transmitting broadcast signals and method for receiving broadcast signals

ABSTRACT

A method and an apparatus for transmitting broadcast signals thereof are disclosed. The apparatus for transmitting broadcast signals comprises an encoder for encoding service data corresponding to each of a plurality of transmission unit, wherein the each of a plurality of transmission unit carries at least one service component, a frame builder for building at least one signal frame including the encoded service data, a phase distortion unit for performing a phase distortion on data in the built at least one signal frame, a modulator for modulating data in the built at least one signal frame by an OFDM (Orthogonal Frequency Division Multiplex) scheme and a transmitter for transmitting the broadcast signals including the modulated data.

This application claims the benefit of earlier filing date and right of priority to U.S. Provisional Application No. 61/873,838, filed on Sep. 5, 2013 and U.S. Provisional Application No. 61/876,734, filed on Sep. 11, 2013 the contents of which are hereby incorporated by reference as if fully set forth herein in their entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to an apparatus for transmitting broadcast signals, an apparatus for receiving broadcast signals and methods for transmitting and receiving broadcast signals.

2. Discussion of the Related Art

As analog broadcast signal transmission comes to an end, various technologies for transmitting/receiving digital broadcast signals are being developed. A digital broadcast signal may include a larger amount of video/audio data than an analog broadcast signal and further include various types of additional data in addition to the video/audio data.

That is, a digital broadcast system can provide HD (high definition) images, multi-channel audio and various additional services. However, data transmission efficiency for transmission of large amounts of data, robustness of transmission/reception networks and network flexibility in consideration of mobile reception equipment need to be improved for digital broadcast.

SUMMARY OF THE INVENTION

Accordingly, the present invention is directed to an apparatus for transmitting broadcast signals and an apparatus for receiving broadcast signals for future broadcast services and methods for transmitting and receiving broadcast signals for future broadcast services.

An object of the present invention is to provide an apparatus and method for transmitting broadcast signals to multiplex data of a broadcast transmission/reception system providing two or more different broadcast services in a time domain and transmit the multiplexed data through the same RF signal bandwidth and an apparatus and method for receiving broadcast signals corresponding thereto.

Another object of the present invention is to provide an apparatus for transmitting broadcast signals, an apparatus for receiving broadcast signals and methods for transmitting and receiving broadcast signals to classify data corresponding to services by components, transmit data corresponding to each component as a data pipe, receive and process the data

Still another object of the present invention is to provide an apparatus for transmitting broadcast signals, an apparatus for receiving broadcast signals and methods for transmitting and receiving broadcast signals to signal signaling information necessary to provide broadcast signals.

Technical Solution

To achieve the object and other advantages and in accordance with the purpose of the invention, as embodied and broadly described herein, a method for transmitting broadcast signals comprises encoding service data corresponding to each of a plurality of transmission unit, wherein the each of a plurality of transmission unit carries at least one service component, building at least one signal frame including the encoded service data, performing a phase distortion on data in the built at least one signal frame, modulating data in the built at least one signal frame by an OFDM (Orthogonal Frequency Division Multiplex) scheme and transmitting the broadcast signals including the modulated data.

Advantageous Effects

The present invention can process data according to service characteristics to control QoS for each service or service component, thereby providing various broadcast services.

The present invention can achieve transmission flexibility by transmitting various broadcast services through the same RF signal bandwidth.

The present invention can improve data transmission efficiency and increase robustness of transmission/reception of broadcast signals using a MIMO system.

According to the present invention, it is possible to provide broadcast signal transmission and reception methods and apparatus capable of receiving digital broadcast signals without error even with mobile reception equipment or in an indoor environment.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates a structure of an apparatus for transmitting broadcast signals for future broadcast services according to an embodiment of the present invention.

FIG. 2 illustrates an input formatting module according to an embodiment of the present invention.

FIG. 3 illustrates an input formatting module according to another embodiment of the present invention.

FIG. 4 illustrates an input formatting module according to another embodiment of the present invention.

FIG. 5 illustrates a coding & modulation module according to an embodiment of the present invention.

FIG. 6 illustrates a frame structure module according to an embodiment of the present invention.

FIG. 7 illustrates a waveform generation module according to an embodiment of the present invention.

FIG. 8 illustrates a structure of an apparatus for receiving broadcast signals for future broadcast services according to an embodiment of the present invention.

FIG. 9 illustrates a synchronization & demodulation module according to an embodiment of the present invention.

FIG. 10 illustrates a frame parsing module according to an embodiment of the present invention.

FIG. 11 illustrates a demapping & decoding module according to an embodiment of the present invention.

FIG. 12 illustrates an output processor according to an embodiment of the present invention.

FIG. 13 illustrates an output processor according to another embodiment of the present invention.

FIG. 14 illustrates a coding & modulation module according to another embodiment of the present invention.

FIG. 15 illustrates a demapping & decoding module according to another embodiment of the present invention.

FIG. 16 is a view illustrating a waveform generation module according to another embodiment of the present invention.

FIG. 17 is graphs related to the phase transition according to the first scheme of the present invention.

FIG. 18 is a conceptual diagram of phase transition according to the second scheme of the present invention.

FIG. 19 illustrates a phase transition operation according to the third scheme of the present invention.

FIG. 20 is a conceptual diagram of phase transition in the frequency dimension according to a conventional technology.

FIG. 21 is a conceptual diagram illustrating phase transition according to embodiments of the present invention.

FIG. 22 illustrates phase transition according to the first scheme of the present invention.

FIG. 23 illustrates another example the conceptual diagram of phase transition according to the second scheme of the present invention.

FIG. 24 illustrates another example of the phase transition operation according to the third scheme of the present invention.

FIG. 25 illustrates a waveform generation module according to another example of the present invention.

FIG. 26 is an equation indicating 2D phase pre-distortion according to embodiments of the present invention.

FIG. 27 illustrates a phase transition operation according to the third scheme of the 2D phase pre-distortion according to embodiments of the present invention.

FIG. 28 is a flowchart illustrating a method for transmitting broadcast signals according to an embodiment of the present invention.

FIG. 29 is a flowchart illustrating a method for receiving broadcast signals according to an embodiment of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

Reference will now be made in detail to the preferred embodiments of the present invention, examples of which are illustrated in the accompanying drawings. The detailed description, which will be given below with reference to the accompanying drawings, is intended to explain exemplary embodiments of the present invention, rather than to show the only embodiments that can be implemented according to the present invention. The following detailed description includes specific details in order to provide a thorough understanding of the present invention. However, it will be apparent to those skilled in the art that the present invention may be practiced without such specific details.

Although most terms used in the present invention have been selected from general ones widely used in the art, some terms have been arbitrarily selected by the applicant and their meanings are explained in detail in the following description as needed. Thus, the present invention should be understood based upon the intended meanings of the terms rather than their simple names or meanings.

The present invention provides apparatuses and methods for transmitting and receiving broadcast signals for future broadcast services. Future broadcast services according to an embodiment of the present invention include a terrestrial broadcast service, a mobile broadcast service, a UHDTV service, etc. The apparatuses and methods for transmitting according to an embodiment of the present invention may be categorized into a base profile for the terrestrial broadcast service, a handheld profile for the mobile broadcast service and an advanced profile for the UHDTV service. In this case, the base profile can be used as a profile for both the terrestrial broadcast service and the mobile broadcast service. That is, the base profile can be used to define a concept of a profile which includes the mobile profile. This can be changed according to intention of the designer.

The present invention may process broadcast signals for the future broadcast services through non-MIMO (Multiple Input Multiple Output) or MIMO according to one embodiment. A non-MIMO scheme according to an embodiment of the present invention may include a MISO (Multiple Input Single Output) scheme, a SISO (Single Input Single Output) scheme, etc.

While MISO or MIMO uses two antennas in the following for convenience of description, the present invention is applicable to systems using two or more antennas.

FIG. 1 illustrates a structure of an apparatus for transmitting broadcast signals for future broadcast services according to an embodiment of the present invention.

The apparatus for transmitting broadcast signals for future broadcast services according to an embodiment of the present invention can include an input formatting module 1000, a coding & modulation module 1100, a frame structure module 1200, a waveform generation module 1300 and a signaling generation module 1400. A description will be given of the operation of each module of the apparatus for transmitting broadcast signals.

Referring to FIG. 1, the apparatus for transmitting broadcast signals for future broadcast services according to an embodiment of the present invention can receive MPEG-TSs, IP streams (v4/v6) and generic streams (GSs) as an input signal. In addition, the apparatus for transmitting broadcast signals can receive management information about the configuration of each stream constituting the input signal and generate a final physical layer signal with reference to the received management information.

The input formatting module 1000 according to an embodiment of the present invention can classify the input streams on the basis of a standard for coding and modulation or services or service components and output the input streams as a plurality of logical data pipes (or data pipes or DP data). The data pipe is a logical channel in the physical layer that carries service data or related metadata, which may carry one or multiple service(s) or service component(s). In addition, data transmitted through each data pipe may be called DP data.

In addition, the input formatting module 1000 according to an embodiment of the present invention can divide each data pipe into blocks necessary to perform coding and modulation and carry out processes necessary to increase transmission efficiency or to perform scheduling. Details of operations of the input formatting module 1000 will be described later.

The coding & modulation module 1100 according to an embodiment of the present invention can perform forward error correction (FEC) encoding on each data pipe received from the input formatting module 1000 such that an apparatus for receiving broadcast signals can correct an error that may be generated on a transmission channel. In addition, the coding & modulation module 1100 according to an embodiment of the present invention can convert FEC output bit data to symbol data and interleave the symbol data to correct burst error caused by a channel. As shown in FIG. 1, the coding & modulation module 1100 according to an embodiment of the present invention can divide the processed data such that the divided data can be output through data paths for respective antenna outputs in order to transmit the data through two or more Tx antennas.

The frame structure module 1200 according to an embodiment of the present invention can map the data output from the coding & modulation module 1100 to signal frames. The frame structure module 1200 according to an embodiment of the present invention can perform mapping using scheduling information output from the input formatting module 1000 and interleave data in the signal frames in order to obtain additional diversity gain.

The waveform generation module 1300 according to an embodiment of the present invention can convert the signal frames output from the frame structure module 1200 into a signal for transmission. In this case, the waveform generation module 1300 according to an embodiment of the present invention can insert a preamble signal (or preamble) into the signal for detection of the transmission apparatus and insert a reference signal for estimating a transmission channel to compensate for distortion into the signal. In addition, the waveform generation module 1300 according to an embodiment of the present invention can provide a guard interval and insert a specific sequence into the same in order to offset the influence of channel delay spread due to multi-path reception. Additionally, the waveform generation module 1300 according to an embodiment of the present invention can perform a procedure necessary for efficient transmission in consideration of signal characteristics such as a peak-to-average power ratio of the output signal.

The signaling generation module 1400 according to an embodiment of the present invention generates final physical layer signaling information using the input management information and information generated by the input formatting module 1000, coding & modulation module 1100 and frame structure module 1200. Accordingly, a reception apparatus according to an embodiment of the present invention can decode a received signal by decoding the signaling information.

As described above, the apparatus for transmitting broadcast signals for future broadcast services according to one embodiment of the present invention can provide terrestrial broadcast service, mobile broadcast service, UHDTV service, etc. Accordingly, the apparatus for transmitting broadcast signals for future broadcast services according to one embodiment of the present invention can multiplex signals for different services in the time domain and transmit the same.

FIGS. 2, 3 and 4 illustrate the input formatting module 1000 according to embodiments of the present invention. A description will be given of each figure.

FIG. 2 illustrates an input formatting module according to one embodiment of the present invention. FIG. 2 shows an input formatting module when the input signal is a single input stream.

Referring to FIG. 2, the input formatting module according to one embodiment of the present invention can include a mode adaptation module 2000 and a stream adaptation module 2100.

As shown in FIG. 2, the mode adaptation module 2000 can include an input interface block 2010, a CRC-8 encoder block 2020 and a BB header insertion block 2030. Description will be given of each block of the mode adaptation module 2000.

The input interface block 2010 can divide the single input stream input thereto into data pieces each having the length of a baseband (BB) frame used for FEC (BCH/LDPC) which will be performed later and output the data pieces.

The CRC-8 encoder block 2020 can perform CRC encoding on BB frame data to add redundancy data thereto.

The BB header insertion block 2030 can insert, into the BB frame data, a header including information such as mode adaptation type (TS/GS/IP), a user packet length, a data field length, user packet sync byte, start address of user packet sync byte in data field, a high efficiency mode indicator, an input stream synchronization field, etc.

As shown in FIG. 2, the stream adaptation module 2100 can include a padding insertion block 2110 and a BB scrambler block 2120. Description will be given of each block of the stream adaptation module 2100.

If data received from the mode adaptation module 2000 has a length shorter than an input data length necessary for FEC encoding, the padding insertion block 2110 can insert a padding bit into the data such that the data has the input data length and output the data including the padding bit.

The BB scrambler block 2120 can randomize the input bit stream by performing an XOR operation on the input bit stream and a pseudo random binary sequence (PRBS).

The above-described blocks may be omitted or replaced by blocks having similar or identical functions.

As shown in FIG. 2, the input formatting module can finally output data pipes to the coding & modulation module.

FIG. 3 illustrates an input formatting module according to another embodiment of the present invention. FIG. 3 shows a mode adaptation module 3000 of the input formatting module when the input signal corresponds to multiple input streams.

The mode adaptation module 3000 of the input formatting module for processing the multiple input streams can independently process the multiple input streams.

Referring to FIG. 3, the mode adaptation module 3000 for respectively processing the multiple input streams can include input interface blocks, input stream synchronizer blocks 3100, compensating delay blocks 3200, null packet deletion blocks 3300, CRC-8 encoder blocks and BB header insertion blocks. Description will be given of each block of the mode adaptation module 3000.

Operations of the input interface block, CRC-8 encoder block and BB header insertion block correspond to those of the input interface block, CRC-8 encoder block and BB header insertion block described with reference to FIG. 2 and thus description thereof is omitted.

The input stream synchronizer block 3100 can transmit input stream clock reference (ISCR) information to generate timing information necessary for the apparatus for receiving broadcast signals to restore the TSs or GSs.

The compensating delay block 3200 can delay input data and output the delayed input data such that the apparatus for receiving broadcast signals can synchronize the input data if a delay is generated between data pipes according to processing of data including the timing information by the transmission apparatus.

The null packet deletion block 3300 can delete unnecessarily transmitted input null packets from the input data, insert the number of deleted null packets into the input data based on positions in which the null packets are deleted and transmit the input data.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions.

FIG. 4 illustrates an input formatting module according to another embodiment of the present invention.

Specifically, FIG. 4 illustrates a stream adaptation module of the input formatting module when the input signal corresponds to multiple input streams.

The stream adaptation module of the input formatting module when the input signal corresponds to multiple input streams can include a scheduler 4000, a 1-frame delay block 4100, an in-band signaling or padding insertion block 4200, a physical layer signaling generation block 4300 and a BB scrambler block 4400. Description will be given of each block of the stream adaptation module.

The scheduler 4000 can perform scheduling for a MIMO system using multiple antennas having dual polarity. In addition, the scheduler 4000 can generate parameters for use in signal processing blocks for antenna paths, such as a bit-to-cell demux block, a cell interleaver block, a time interleaver block, etc. included in the coding & modulation module illustrated in FIG. 1.

The 1-frame delay block 4100 can delay the input data by one transmission frame such that scheduling information about the next frame can be transmitted through the current frame for in-band signaling information to be inserted into the data pipes.

The in-band signaling or padding insertion block 4200 can insert undelayed physical layer signaling (PLS)-dynamic signaling information into the data delayed by one transmission frame. In this case, the in-band signaling or padding insertion block 4200 can insert a padding bit when a space for padding is present or insert in-band signaling information into the padding space. In addition, the scheduler 4000 can output physical layer signaling-dynamic signaling information about the current frame separately from in-band signaling information. Accordingly, a cell mapper, which will be described later, can map input cells according to scheduling information output from the scheduler 4000.

The physical layer signaling generation block 4300 can generate physical layer signaling data which will be transmitted through a preamble symbol of a transmission frame or spread and transmitted through a data symbol other than the in-band signaling information. In this case, the physical layer signaling data according to an embodiment of the present invention can be referred to as signaling information. Furthermore, the physical layer signaling data according to an embodiment of the present invention can be divided into PLS-pre information and PLS-post information. The PLS-pre information can include parameters necessary to encode the PLS-post information and static PLS signaling data and the PLS-post information can include parameters necessary to encode the data pipes. The parameters necessary to encode the data pipes can be classified into static PLS signaling data and dynamic PLS signaling data. The static PLS signaling data is a parameter commonly applicable to all frames included in a super-frame and can be changed on a super-frame basis. The dynamic PLS signaling data is a parameter differently applicable to respective frames included in a super-frame and can be changed on a frame-by-frame basis. Accordingly, the reception apparatus can acquire the PLS-post information by decoding the PLS-pre information and decode desired data pipes by decoding the PLS-post information.

The BB scrambler block 4400 can generate a pseudo-random binary sequence (PRBS) and perform an XOR operation on the PRBS and the input bit streams to decrease the peak-to-average power ratio (PAPR) of the output signal of the waveform generation block. As shown in FIG. 4, scrambling of the BB scrambler block 4400 is applicable to both data pipes and physical layer signaling information.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions according to designer.

As shown in FIG. 4, the stream adaptation module can finally output the data pipes to the coding & modulation module.

FIG. 5 illustrates a coding & modulation module according to an embodiment of the present invention.

The coding & modulation module shown in FIG. 5 corresponds to an embodiment of the coding & modulation module illustrated in FIG. 1.

As described above, the apparatus for transmitting broadcast signals for future broadcast services according to an embodiment of the present invention can provide a terrestrial broadcast service, mobile broadcast service, UHDTV service, etc.

Since QoS (quality of service) depends on characteristics of a service provided by the apparatus for transmitting broadcast signals for future broadcast services according to an embodiment of the present invention, data corresponding to respective services needs to be processed through different schemes. Accordingly, the coding & modulation module according to an embodiment of the present invention can independently process data pipes input thereto by independently applying SISO, MISO and MIMO schemes to the data pipes respectively corresponding to data paths. Consequently, the apparatus for transmitting broadcast signals for future broadcast services according to an embodiment of the present invention can control QoS for each service or service component transmitted through each data pipe.

Accordingly, the coding & modulation module according to an embodiment of the present invention can include a first block 5000 for SISO, a second block 5100 for MISO, a third block 5200 for MIMO and a fourth block 5300 for processing the PLS-pre/PLS-post information. The coding & modulation module illustrated in FIG. 5 is an exemplary and may include only the first block 5000 and the fourth block 5300, the second block 5100 and the fourth block 5300 or the third block 5200 and the fourth block 5300 according to design. That is, the coding & modulation module can include blocks for processing data pipes equally or differently according to design.

A description will be given of each block of the coding & modulation module.

The first block 5000 processes an input data pipe according to SISO and can include an FEC encoder block 5010, a bit interleaver block 5020, a bit-to-cell demux block 5030, a constellation mapper block 5040, a cell interleaver block 5050 and a time interleaver block 5060.

The FEC encoder block 5010 can perform BCH encoding and LDPC encoding on the input data pipe to add redundancy thereto such that the reception apparatus can correct an error generated on a transmission channel.

The bit interleaver block 5020 can interleave bit streams of the FEC-encoded data pipe according to an interleaving rule such that the bit streams have robustness against burst error that may be generated on the transmission channel. Accordingly, when deep fading or erasure is applied to QAM symbols, errors can be prevented from being generated in consecutive bits from among all codeword bits since interleaved bits are mapped to the QAM symbols.

The bit-to-cell demux block 5030 can determine the order of input bit streams such that each bit in an FEC block can be transmitted with appropriate robustness in consideration of both the order of input bit streams and a constellation mapping rule.

In addition, the bit interleaver block 5020 is located between the FEC encoder block 5010 and the constellation mapper block 5040 and can connect output bits of LDPC encoding performed by the FEC encoder block 5010 to bit positions having different reliability values and optimal values of the constellation mapper in consideration of LDPC decoding of the apparatus for receiving broadcast signals. Accordingly, the bit-to-cell demux block 5030 can be replaced by a block having a similar or equal function.

The constellation mapper block 5040 can map a bit word input thereto to one constellation. In this case, the constellation mapper block 5040 can additionally perform rotation & Q-delay. That is, the constellation mapper block 5040 can rotate input constellations according to a rotation angle, divide the constellations into an in-phase component and a quadrature-phase component and delay only the quadrature-phase component by an arbitrary value. Then, the constellation mapper block 5040 can remap the constellations to new constellations using a paired in-phase component and quadrature-phase component.

In addition, the constellation mapper block 5040 can move constellation points on a two-dimensional plane in order to find optimal constellation points. Through this process, capacity of the coding & modulation module 1100 can be optimized. Furthermore, the constellation mapper block 5040 can perform the above-described operation using IQ-balanced constellation points and rotation. The constellation mapper block 5040 can be replaced by a block having a similar or equal function.

The cell interleaver block 5050 can randomly interleave cells corresponding to one FEC block and output the interleaved cells such that cells corresponding to respective FEC blocks can be output in different orders.

The time interleaver block 5060 can interleave cells belonging to a plurality of FEC blocks and output the interleaved cells. Accordingly, the cells corresponding to the FEC blocks are dispersed and transmitted in a period corresponding to a time interleaving depth and thus diversity gain can be obtained.

The second block 5100 processes an input data pipe according to MISO and can include the FEC encoder block, bit interleaver block, bit-to-cell demux block, constellation mapper block, cell interleaver block and time interleaver block in the same manner as the first block 5000. However, the second block 5100 is distinguished from the first block 5000 in that the second block 5100 further includes a MISO processing block 5110. The second block 5100 performs the same procedure including the input operation to the time interleaver operation as those of the first block 5000 and thus description of the corresponding blocks is omitted.

The MISO processing block 5110 can encode input cells according to a MISO encoding matrix providing transmit diversity and output MISO-processed data through two paths. MISO processing according to one embodiment of the present invention can include OSTBC (orthogonal space time block coding)/OSFBC (orthogonal space frequency block coding, Alamouti coding).

The third block 5200 processes an input data pipe according to MIMO and can include the FEC encoder block, bit interleaver block, bit-to-cell demux block, constellation mapper block, cell interleaver block and time interleaver block in the same manner as the second block 5100, as shown in FIG. 5. However, the data processing procedure of the third block 5200 is different from that of the second block 5100 since the third block 5200 includes a MIMO processing block 5220.

That is, in the third block 5200, basic roles of the FEC encoder block and the bit interleaver block are identical to those of the first and second blocks 5000 and 5100 although functions thereof may be different from those of the first and second blocks 5000 and 5100.

The bit-to-cell demux block 5210 can generate as many output bit streams as input bit streams of MIMO processing and output the output bit streams through MIMO paths for MIMO processing. In this case, the bit-to-cell demux block 5210 can be designed to optimize the decoding performance of the reception apparatus in consideration of characteristics of LDPC and MIMO processing.

Basic roles of the constellation mapper block, cell interleaver block and time interleaver block are identical to those of the first and second blocks 5000 and 5100 although functions thereof may be different from those of the first and second blocks 5000 and 5100. As shown in FIG. 5, as many constellation mapper blocks, cell interleaver blocks and time interleaver blocks as the number of MIMO paths for MIMO processing can be present. In this case, the constellation mapper blocks, cell interleaver blocks and time interleaver blocks can operate equally or independently for data input through the respective paths.

The MIMO processing block 5220 can perform MIMO processing on two input cells using a MIMO encoding matrix and output the MIMO-processed data through two paths. The MIMO encoding matrix according to an embodiment of the present invention can include spatial multiplexing, Golden code, full-rate full diversity code, linear dispersion code, etc.

The fourth block 5300 processes the PLS-pre/PLS-post information and can perform SISO or MISO processing.

The basic roles of the bit interleaver block, bit-to-cell demux block, constellation mapper block, cell interleaver block, time interleaver block and MISO processing block included in the fourth block 5300 correspond to those of the second block 5100 although functions thereof may be different from those of the second block 5100.

A shortened/punctured FEC encoder block 5310 included in the fourth block 5300 can process PLS data using an FEC encoding scheme for a PLS path provided for a case in which the length of input data is shorter than a length necessary to perform FEC encoding. Specifically, the shortened/punctured FEC encoder block 5310 can perform BCH encoding on input bit streams, pad 0s corresponding to a desired input bit stream length necessary for normal LDPC encoding, carry out LDPC encoding and then remove the padded 0s to puncture parity bits such that an effective code rate becomes equal to or lower than the data pipe rate.

The blocks included in the first block 5000 to fourth block 5300 may be omitted or replaced by blocks having similar or identical functions according to design.

As illustrated in FIG. 5, the coding & modulation module can output the data pipes (or DP data), PLS-pre information and PLS-post information processed for the respective paths to the frame structure module.

FIG. 6 illustrates a frame structure module according to one embodiment of the present invention.

The frame structure module shown in FIG. 6 corresponds to an embodiment of the frame structure module 1200 illustrated in FIG. 1.

The frame structure module according to one embodiment of the present invention can include at least one cell-mapper 6000, at least one delay compensation module 6100 and at least one block interleaver 6200. The number of cell mappers 6000, delay compensation modules 6100 and block interleavers 6200 can be changed. A description will be given of each module of the frame structure block.

The cell-mapper 6000 can allocate cells corresponding to SISO-, MISO- or MIMO-processed data pipes output from the coding & modulation module, cells corresponding to common data commonly applicable to the data pipes and cells corresponding to the PLS-pre/PLS-post information to signal frames according to scheduling information. The common data refers to signaling information commonly applied to all or some data pipes and can be transmitted through a specific data pipe. The data pipe through which the common data is transmitted can be referred to as a common data pipe and can be changed according to design.

When the apparatus for transmitting broadcast signals according to an embodiment of the present invention uses two output antennas and Alamouti coding is used for MISO processing, the cell-mapper 6000 can perform pair-wise cell mapping in order to maintain orthogonality according to Alamouti encoding. That is, the cell-mapper 6000 can process two consecutive cells of the input cells as one unit and map the unit to a frame. Accordingly, paired cells in an input path corresponding to an output path of each antenna can be allocated to neighboring positions in a transmission frame.

The delay compensation block 6100 can obtain PLS data corresponding to the current transmission frame by delaying input PLS data cells for the next transmission frame by one frame. In this case, the PLS data corresponding to the current frame can be transmitted through a preamble part in the current signal frame and PLS data corresponding to the next signal frame can be transmitted through a preamble part in the current signal frame or in-band signaling in each data pipe of the current signal frame. This can be changed by the designer.

The block interleaver 6200 can obtain additional diversity gain by interleaving cells in a transport block corresponding to the unit of a signal frame. In addition, the block interleaver 6200 can perform interleaving by processing two consecutive cells of the input cells as one unit when the above-described pair-wise cell mapping is performed. Accordingly, cells output from the block interleaver 6200 can be two consecutive identical cells.

When pair-wise mapping and pair-wise interleaving are performed, at least one cell mapper and at least one block interleaver can operate equally or independently for data input through the paths.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions according to design.

As illustrated in FIG. 6, the frame structure module can output at least one signal frame to the waveform generation module.

FIG. 7 illustrates a waveform generation module according to an embodiment of the present invention.

The waveform generation module illustrated in FIG. 7 corresponds to an embodiment of the waveform generation module 1300 described with reference to FIG. 1.

The waveform generation module according to an embodiment of the present invention can modulate and transmit as many signal frames as the number of antennas for receiving and outputting signal frames output from the frame structure module illustrated in FIG. 6.

Specifically, the waveform generation module illustrated in FIG. 7 is an embodiment of a waveform generation module of an apparatus for transmitting broadcast signals using m Tx antennas and can include m processing blocks for modulating and outputting frames corresponding to m paths. The m processing blocks can perform the same processing procedure. A description will be given of operation of the first processing block 7000 from among the m processing blocks.

The first processing block 7000 can include a reference signal & PAPR reduction block 7100, an inverse waveform transform block 7200, a PAPR reduction in time block 7300, a guard sequence insertion block 7400, a preamble insertion block 7500, a waveform processing block 7600, other system insertion block 7700 and a DAC (digital analog converter) block 7800.

The reference signal insertion & PAPR reduction block 7100 can insert a reference signal into a predetermined position of each signal block and apply a PAPR reduction scheme to reduce a PAPR in the time domain. If a broadcast transmission/reception system according to an embodiment of the present invention corresponds to an OFDM system, the reference signal insertion & PAPR reduction block 7100 can use a method of reserving some active subcarriers rather than using the same. In addition, the reference signal insertion & PAPR reduction block 7100 may not use the PAPR reduction scheme as an optional feature according to broadcast transmission/reception system.

The inverse waveform transform block 7200 can transform an input signal in a manner of improving transmission efficiency and flexibility in consideration of transmission channel characteristics and system architecture. If the broadcast transmission/reception system according to an embodiment of the present invention corresponds to an OFDM system, the inverse waveform transform block 7200 can employ a method of transforming a frequency domain signal into a time domain signal through inverse FFT operation. If the broadcast transmission/reception system according to an embodiment of the present invention corresponds to a single carrier system, the inverse waveform transform block 7200 may not be used in the waveform generation module.

The PAPR reduction in time block 7300 can use a method for reducing PAPR of an input signal in the time domain. If the broadcast transmission/reception system according to an embodiment of the present invention corresponds to an OFDM system, the PAPR reduction in time block 7300 may use a method of simply clipping peak amplitude. Furthermore, the PAPR reduction in time block 7300 may not be used in the broadcast transmission/reception system according to an embodiment of the present invention since it is an optional feature.

The guard sequence insertion block 7400 can provide a guard interval between neighboring signal blocks and insert a specific sequence into the guard interval as necessary in order to minimize the influence of delay spread of a transmission channel. Accordingly, the reception apparatus can easily perform synchronization or channel estimation. If the broadcast transmission/reception system according to an embodiment of the present invention corresponds to an OFDM system, the guard sequence insertion block 7400 may insert a cyclic prefix into a guard interval of an OFDM symbol.

The preamble insertion block 7500 can insert a signal of a known type (e.g. the preamble or preamble symbol) agreed upon between the transmission apparatus and the reception apparatus into a transmission signal such that the reception apparatus can rapidly and efficiently detect a target system signal. If the broadcast transmission/reception system according to an embodiment of the present invention corresponds to an OFDM system, the preamble insertion block 7500 can define a signal frame composed of a plurality of OFDM symbols and insert a preamble symbol into the beginning of each signal frame. That is, the preamble carries basic PLS data and is located in the beginning of a signal frame.

The waveform processing block 7600 can perform waveform processing on an input baseband signal such that the input baseband signal meets channel transmission characteristics. The waveform processing block 7600 may use a method of performing square-root-raised cosine (SRRC) filtering to obtain a standard for out-of-band emission of a transmission signal. If the broadcast transmission/reception system according to an embodiment of the present invention corresponds to a multi-carrier system, the waveform processing block 7600 may not be used.

The other system insertion block 7700 can multiplex signals of a plurality of broadcast transmission/reception systems in the time domain such that data of two or more different broadcast transmission/reception systems providing broadcast services can be simultaneously transmitted in the same RF signal bandwidth. In this case, the two or more different broadcast transmission/reception systems refer to systems providing different broadcast services. The different broadcast services may refer to a terrestrial broadcast service, mobile broadcast service, etc. Data related to respective broadcast services can be transmitted through different frames.

The DAC block 7800 can convert an input digital signal into an analog signal and output the analog signal. The signal output from the DAC block 7800 can be transmitted through m output antennas. A Tx antenna according to an embodiment of the present invention can have vertical or horizontal polarity.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions according to design.

FIG. 8 illustrates a structure of an apparatus for receiving broadcast signals for future broadcast services according to an embodiment of the present invention.

The apparatus for receiving broadcast signals for future broadcast services according to an embodiment of the present invention can correspond to the apparatus for transmitting broadcast signals for future broadcast services, described with reference to FIG. 1. The apparatus for receiving broadcast signals for future broadcast services according to an embodiment of the present invention can include a synchronization & demodulation module 8000, a frame parsing module 8100, a demapping & decoding module 8200, an output processor 8300 and a signaling decoding module 8400. A description will be given of operation of each module of the apparatus for receiving broadcast signals.

The synchronization & demodulation module 8000 can receive input signals through m Rx antennas, perform signal detection and synchronization with respect to a system corresponding to the apparatus for receiving broadcast signals and carry out demodulation corresponding to a reverse procedure of the procedure performed by the apparatus for transmitting broadcast signals.

The frame parsing module 8100 can parse input signal frames and extract data through which a service selected by a user is transmitted. If the apparatus for transmitting broadcast signals performs interleaving, the frame parsing module 8100 can carry out deinterleaving corresponding to a reverse procedure of interleaving. In this case, the positions of a signal and data that need to be extracted can be obtained by decoding data output from the signaling decoding module 8400 to restore scheduling information generated by the apparatus for transmitting broadcast signals.

The demapping & decoding module 8200 can convert the input signals into bit domain data and then deinterleave the same as necessary. The demapping & decoding module 8200 can perform demapping for mapping applied for transmission efficiency and correct an error generated on a transmission channel through decoding. In this case, the demapping & decoding module 8200 can obtain transmission parameters necessary for demapping and decoding by decoding the data output from the signaling decoding module 8400.

The output processor 8300 can perform reverse procedures of various compression/signal processing procedures which are applied by the apparatus for transmitting broadcast signals to improve transmission efficiency. In this case, the output processor 8300 can acquire necessary control information from data output from the signaling decoding module 8400. The output of the output processor 8300 corresponds to a signal input to the apparatus for transmitting broadcast signals and may be MPEG-TSs, IP streams (v4 or v6) and generic streams.

The signaling decoding module 8400 can obtain PLS information from the signal demodulated by the synchronization & demodulation module 8000. As described above, the frame parsing module 8100, demapping & decoding module 8200 and output processor 8300 can execute functions thereof using the data output from the signaling decoding module 8400.

FIG. 9 illustrates a synchronization & demodulation module according to an embodiment of the present invention.

The synchronization & demodulation module shown in FIG. 9 corresponds to an embodiment of the synchronization & demodulation module described with reference to FIG. 8. The synchronization & demodulation module shown in FIG. 9 can perform a reverse operation of the operation of the waveform generation module illustrated in FIG. 7.

As shown in FIG. 9, the synchronization & demodulation module according to an embodiment of the present invention corresponds to a synchronization & demodulation module of an apparatus for receiving broadcast signals using m Rx antennas and can include m processing blocks for demodulating signals respectively input through m paths. The m processing blocks can perform the same processing procedure. A description will be given of operation of the first processing block 9000 from among the m processing blocks.

The first processing block 9000 can include a tuner 9100, an ADC block 9200, a preamble detector 9300, a guard sequence detector 9400, a waveform transform block 9500, a time/frequency synchronization block 9600, a reference signal detector 9700, a channel equalizer 9800 and an inverse waveform transform block 9900.

The tuner 9100 can select a desired frequency band, compensate for the magnitude of a received signal and output the compensated signal to the ADC block 9200.

The ADC block 9200 can convert the signal output from the tuner 9100 into a digital signal.

The preamble detector 9300 can detect a preamble (or preamble signal or preamble symbol) in order to check whether or not the digital signal is a signal of the system corresponding to the apparatus for receiving broadcast signals. In this case, the preamble detector 9300 can decode basic transmission parameters received through the preamble.

The guard sequence detector 9400 can detect a guard sequence in the digital signal. The time/frequency synchronization block 9600 can perform time/frequency synchronization using the detected guard sequence and the channel equalizer 9800 can estimate a channel through a received/restored sequence using the detected guard sequence.

The waveform transform block 9500 can perform a reverse operation of inverse waveform transform when the apparatus for transmitting broadcast signals has performed inverse waveform transform. When the broadcast transmission/reception system according to one embodiment of the present invention is a multi-carrier system, the waveform transform block 9500 can perform FFT. Furthermore, when the broadcast transmission/reception system according to an embodiment of the present invention is a single carrier system, the waveform transform block 9500 may not be used if a received time domain signal is processed in the frequency domain or processed in the time domain.

The time/frequency synchronization block 9600 can receive output data of the preamble detector 9300, guard sequence detector 9400 and reference signal detector 9700 and perform time synchronization and carrier frequency synchronization including guard sequence detection and block window positioning on a detected signal. Here, the time/frequency synchronization block 9600 can feed back the output signal of the waveform transform block 9500 for frequency synchronization.

The reference signal detector 9700 can detect a received reference signal. Accordingly, the apparatus for receiving broadcast signals according to an embodiment of the present invention can perform synchronization or channel estimation.

The channel equalizer 9800 can estimate a transmission channel from each Tx antenna to each Rx antenna from the guard sequence or reference signal and perform channel equalization for received data using the estimated channel.

The inverse waveform transform block 9900 may restore the original received data domain when the waveform transform block 9500 performs waveform transform for efficient synchronization and channel estimation/equalization. If the broadcast transmission/reception system according to an embodiment of the present invention is a single carrier system, the waveform transform block 9500 can perform FFT in order to carry out synchronization/channel estimation/equalization in the frequency domain and the inverse waveform transform block 9900 can perform IFFT on the channel-equalized signal to restore transmitted data symbols. If the broadcast transmission/reception system according to an embodiment of the present invention is a multi-carrier system, the inverse waveform transform block 9900 may not be used.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions according to design.

FIG. 10 illustrates a frame parsing module according to an embodiment of the present invention.

The frame parsing module illustrated in FIG. 10 corresponds to an embodiment of the frame parsing module described with reference to FIG. 8. The frame parsing module shown in FIG. 10 can perform a reverse operation of the operation of the frame structure module illustrated in FIG. 6.

As shown in FIG. 10, the frame parsing module according to an embodiment of the present invention can include at least one block deinterleaver 10000 and at least one cell demapper 10100.

The block deinterleaver 10000 can deinterleave data input through data paths of the m Rx antennas and processed by the synchronization & demodulation module on a signal block basis. In this case, if the apparatus for transmitting broadcast signals performs pair-wise interleaving as illustrated in FIG. 8, the block deinterleaver 10000 can process two consecutive pieces of data as a pair for each input path. Accordingly, the block interleaver 10000 can output two consecutive pieces of data even when deinterleaving has been performed. Furthermore, the block deinterleaver 10000 can perform a reverse operation of the interleaving operation performed by the apparatus for transmitting broadcast signals to output data in the original order.

The cell demapper 10100 can extract cells corresponding to common data, cells corresponding to data pipes and cells corresponding to PLS data from received signal frames. The cell demapper 10100 can merge data distributed and transmitted and output the same as a stream as necessary. When two consecutive pieces of cell input data are processed as a pair and mapped in the apparatus for transmitting broadcast signals, as shown in FIG. 6, the cell demapper 10100 can perform pair-wise cell demapping for processing two consecutive input cells as one unit as a reverse procedure of the mapping operation of the apparatus for transmitting broadcast signals.

In addition, the cell demapper 10100 can extract PLS signaling data received through the current frame as PLS-pre & PLS-post data and output the PLS-pre & PLS-post data.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions according to design.

FIG. 11 illustrates a demapping & decoding module according to an embodiment of the present invention.

The demapping & decoding module shown in FIG. 11 corresponds to an embodiment of the demapping & decoding module illustrated in FIG. 8. The demapping & decoding module shown in FIG. 11 can perform a reverse operation of the operation of the coding & modulation module illustrated in FIG. 5.

The coding & modulation module of the apparatus for transmitting broadcast signals according to an embodiment of the present invention can process input data pipes by independently applying SISO, MISO and MIMO thereto for respective paths, as described above. Accordingly, the demapping & decoding module illustrated in FIG. 11 can include blocks for processing data output from the frame parsing module according to SISO, MISO and MIMO in response to the apparatus for transmitting broadcast signals.

As shown in FIG. 11, the demapping & decoding module according to an embodiment of the present invention can include a first block 11000 for SISO, a second block 11100 for MISO, a third block 11200 for MIMO and a fourth block 11300 for processing the PLS-pre/PLS-post information. The demapping & decoding module shown in FIG. 11 is exemplary and may include only the first block 11000 and the fourth block 11300, only the second block 11100 and the fourth block 11300 or only the third block 11200 and the fourth block 11300 according to design. That is, the demapping & decoding module can include blocks for processing data pipes equally or differently according to design.

A description will be given of each block of the demapping & decoding module.

The first block 11000 processes an input data pipe according to SISO and can include a time deinterleaver block 11010, a cell deinterleaver block 11020, a constellation demapper block 11030, a cell-to-bit mux block 11040, a bit deinterleaver block 11050 and an FEC decoder block 11060.

The time deinterleaver block 11010 can perform a reverse process of the process performed by the time interleaver block 5060 illustrated in FIG. 5. That is, the time deinterleaver block 11010 can deinterleave input symbols interleaved in the time domain into original positions thereof.

The cell deinterleaver block 11020 can perform a reverse process of the process performed by the cell interleaver block 5050 illustrated in FIG. 5. That is, the cell deinterleaver block 11020 can deinterleave positions of cells spread in one FEC block into original positions thereof.

The constellation demapper block 11030 can perform a reverse process of the process performed by the constellation mapper block 5040 illustrated in FIG. 5. That is, the constellation demapper block 11030 can demap a symbol domain input signal to bit domain data. In addition, the constellation demapper block 11030 may perform hard decision and output decided bit data. Furthermore, the constellation demapper block 11030 may output a log-likelihood ratio (LLR) of each bit, which corresponds to a soft decision value or probability value. If the apparatus for transmitting broadcast signals applies a rotated constellation in order to obtain additional diversity gain, the constellation demapper block 11030 can perform 2-dimensional LLR demapping corresponding to the rotated constellation. Here, the constellation demapper block 11030 can calculate the LLR such that a delay applied by the apparatus for transmitting broadcast signals to the I or Q component can be compensated.

The cell-to-bit mux block 11040 can perform a reverse process of the process performed by the bit-to-cell demux block 5030 illustrated in FIG. 5. That is, the cell-to-bit mux block 11040 can restore bit data mapped by the bit-to-cell demux block 5030 to the original bit streams.

The bit deinterleaver block 11050 can perform a reverse process of the process performed by the bit interleaver 5020 illustrated in FIG. 5. That is, the bit deinterleaver block 11050 can deinterleave the bit streams output from the cell-to-bit mux block 11040 in the original order.

The FEC decoder block 11060 can perform a reverse process of the process performed by the FEC encoder block 5010 illustrated in FIG. 5. That is, the FEC decoder block 11060 can correct an error generated on a transmission channel by performing LDPC decoding and BCH decoding.

The second block 11100 processes an input data pipe according to MISO and can include the time deinterleaver block, cell deinterleaver block, constellation demapper block, cell-to-bit mux block, bit deinterleaver block and FEC decoder block in the same manner as the first block 11000, as shown in FIG. 11. However, the second block 11100 is distinguished from the first block 11000 in that the second block 11100 further includes a MISO decoding block 11110. The second block 11100 performs the same procedure including time deinterleaving operation to outputting operation as the first block 11000 and thus description of the corresponding blocks is omitted.

The MISO decoding block 11110 can perform a reverse operation of the operation of the MISO processing block 5110 illustrated in FIG. 5. If the broadcast transmission/reception system according to an embodiment of the present invention uses STBC, the MISO decoding block 11110 can perform Alamouti decoding.

The third block 11200 processes an input data pipe according to MIMO and can include the time deinterleaver block, cell deinterleaver block, constellation demapper block, cell-to-bit mux block, bit deinterleaver block and FEC decoder block in the same manner as the second block 11100, as shown in FIG. 11. However, the third block 11200 is distinguished from the second block 11100 in that the third block 11200 further includes a MIMO decoding block 11210. The basic roles of the time deinterleaver block, cell deinterleaver block, constellation demapper block, cell-to-bit mux block and bit deinterleaver block included in the third block 11200 are identical to those of the corresponding blocks included in the first and second blocks 11000 and 11100 although functions thereof may be different from the first and second blocks 11000 and 11100.

The MIMO decoding block 11210 can receive output data of the cell deinterleaver for input signals of the m Rx antennas and perform MIMO decoding as a reverse operation of the operation of the MIMO processing block 5220 illustrated in FIG. 5. The MIMO decoding block 11210 can perform maximum likelihood decoding to obtain optimal decoding performance or carry out sphere decoding with reduced complexity. Otherwise, the MIMO decoding block 11210 can achieve improved decoding performance by performing MMSE detection or carrying out iterative decoding with MMSE detection.

The fourth block 11300 processes the PLS-pre/PLS-post information and can perform SISO or MISO decoding. The fourth block 11300 can carry out a reverse process of the process performed by the fourth block 5300 described with reference to FIG. 5.

The basic roles of the time deinterleaver block, cell deinterleaver block, constellation demapper block, cell-to-bit mux block and bit deinterleaver block included in the fourth block 11300 are identical to those of the corresponding blocks of the first, second and third blocks 11000, 11100 and 11200 although functions thereof may be different from the first, second and third blocks 11000, 11100 and 11200.

The shortened/punctured FEC decoder 11310 included in the fourth block 11300 can perform a reverse process of the process performed by the shortened/punctured FEC encoder block 5310 described with reference to FIG. 5. That is, the shortened/punctured FEC decoder 11310 can perform de-shortening and de-puncturing on data shortened/punctured according to PLS data length and then carry out FEC decoding thereon. In this case, the FEC decoder used for data pipes can also be used for PLS. Accordingly, additional FEC decoder hardware for the PLS only is not needed and thus system design is simplified and efficient coding is achieved.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions according to design.

The demapping & decoding module according to an embodiment of the present invention can output data pipes and PLS information processed for the respective paths to the output processor, as illustrated in FIG. 11.

FIGS. 12 and 13 illustrate output processors according to embodiments of the present invention.

FIG. 12 illustrates an output processor according to an embodiment of the present invention. The output processor illustrated in FIG. 12 corresponds to an embodiment of the output processor illustrated in FIG. 8. The output processor illustrated in FIG. 12 receives a single data pipe output from the demapping & decoding module and outputs a single output stream. The output processor can perform a reverse operation of the operation of the input formatting module illustrated in FIG. 2.

The output processor shown in FIG. 12 can include a BB scrambler block 12000, a padding removal block 12100, a CRC-8 decoder block 12200 and a BB frame processor block 12300.

The BB scrambler block 12000 can descramble an input bit stream by generating the same PRBS as that used in the apparatus for transmitting broadcast signals for the input bit stream and carrying out an XOR operation on the PRBS and the bit stream.

The padding removal block 12100 can remove padding bits inserted by the apparatus for transmitting broadcast signals as necessary.

The CRC-8 decoder block 12200 can check a block error by performing CRC decoding on the bit stream received from the padding removal block 12100.

The BB frame processor block 12300 can decode information transmitted through a BB frame header and restore MPEG-TSs, IP streams (v4 or v6) or generic streams using the decoded information.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions according to design.

FIG. 13 illustrates an output processor according to another embodiment of the present invention. The output processor shown in FIG. 13 corresponds to an embodiment of the output processor illustrated in FIG. 8. The output processor shown in FIG. 13 receives multiple data pipes output from the demapping & decoding module. Decoding multiple data pipes can include a process of merging common data commonly applicable to a plurality of data pipes and data pipes related thereto and decoding the same or a process of simultaneously decoding a plurality of services or service components (including a scalable video service) by the apparatus for receiving broadcast signals.

The output processor shown in FIG. 13 can include a BB descrambler block, a padding removal block, a CRC-8 decoder block and a BB frame processor block as the output processor illustrated in FIG. 12. The basic roles of these blocks correspond to those of the blocks described with reference to FIG. 12 although operations thereof may differ from those of the blocks illustrated in FIG. 12.

A de-jitter buffer block 13000 included in the output processor shown in FIG. 13 can compensate for a delay, inserted by the apparatus for transmitting broadcast signals for synchronization of multiple data pipes, according to a restored TTO (time to output) parameter.

A null packet insertion block 13100 can restore a null packet removed from a stream with reference to a restored DNP (deleted null packet) and output common data.

A TS clock regeneration block 13200 can restore time synchronization of output packets based on ISCR (input stream time reference) information.

A TS recombining block 13300 can recombine the common data and data pipes related thereto, output from the null packet insertion block 13100, to restore the original MPEG-TSs, IP streams (v4 or v6) or generic streams. The TTO, DNT and ISCR information can be obtained through the BB frame header.

An in-band signaling decoding block 13400 can decode and output in-band physical layer signaling information transmitted through a padding bit field in each FEC frame of a data pipe.

The output processor shown in FIG. 13 can BB-descramble the PLS-pre information and PLS-post information respectively input through a PLS-pre path and a PLS-post path and decode the descrambled data to restore the original PLS data. The restored PLS data is delivered to a system controller included in the apparatus for receiving broadcast signals. The system controller can provide parameters necessary for the synchronization & demodulation module, frame parsing module, demapping & decoding module and output processor module of the apparatus for receiving broadcast signals.

The above-described blocks may be omitted or replaced by blocks having similar r identical functions according to design.

FIG. 14 illustrates a coding & modulation module according to another embodiment of the present invention.

The coding & modulation module shown in FIG. 14 corresponds to another embodiment of the coding & modulation module illustrated in FIGS. 1 to 5.

To control QoS for each service or service component transmitted through each data pipe, as described above with reference to FIG. 5, the coding & modulation module shown in FIG. 14 can include a first block 14000 for SISO, a second block 14100 for MISO, a third block 14200 for MIMO and a fourth block 14300 for processing the PLS-pre/PLS-post information. In addition, the coding & modulation module can include blocks for processing data pipes equally or differently according to the design. The first to fourth blocks 14000 to 14300 shown in FIG. 14 are similar to the first to fourth blocks 5000 to 5300 illustrated in FIG. 5.

However, the first to fourth blocks 14000 to 14300 shown in FIG. 14 are distinguished from the first to fourth blocks 5000 to 5300 illustrated in FIG. 5 in that a constellation mapper 14010 included in the first to fourth blocks 14000 to 14300 has a function different from the first to fourth blocks 5000 to 5300 illustrated in FIG. 5, a rotation & I/Q interleaver block 14020 is present between the cell interleaver and the time interleaver of the first to fourth blocks 14000 to 14300 illustrated in FIG. 14 and the third block 14200 for MIMO has a configuration different from the third block 5200 for MIMO illustrated in FIG. 5. The following description focuses on these differences between the first to fourth blocks 14000 to 14300 shown in FIG. 14 and the first to fourth blocks 5000 to 5300 illustrated in FIG. 5.

The constellation mapper block 14010 shown in FIG. 14 can map an input bit word to a complex symbol. However, the constellation mapper block 14010 may not perform constellation rotation, differently from the constellation mapper block shown in FIG. 5. The constellation mapper block 14010 shown in FIG. 14 is commonly applicable to the first, second and third blocks 14000, 14100 and 14200, as described above.

The rotation & I/Q interleaver block 14020 can independently interleave in-phase and quadrature-phase components of each complex symbol of cell-interleaved data output from the cell interleaver and output the in-phase and quadrature-phase components on a symbol-by-symbol basis. The number of number of input data pieces and output data pieces of the rotation & I/Q interleaver block 14020 is two or more which can be changed by the designer. In addition, the rotation & I/Q interleaver block 14020 may not interleave the in-phase component.

The rotation & I/Q interleaver block 14020 is commonly applicable to the first to fourth blocks 14000 to 14300, as described above. In this case, whether or not the rotation & I/Q interleaver block 14020 is applied to the fourth block 14300 for processing the PLS-pre/post information can be signaled through the above-described preamble.

The third block 14200 for MIMO can include a Q-block interleaver block 14210 and a complex symbol generator block 14220, as illustrated in FIG. 14.

The Q-block interleaver block 14210 can permute a parity part of an FEC-encoded FEC block received from the FEC encoder. Accordingly, a parity part of an LDPC H matrix can be made into a cyclic structure like an information part. The Q-block interleaver block 14210 can permute the order of output bit blocks having Q size of the LDPC H matrix and then perform row-column block interleaving to generate final bit streams.

The complex symbol generator block 14220 receives the bit streams output from the Q-block interleaver block 14210, maps the bit streams to complex symbols and outputs the complex symbols. In this case, the complex symbol generator block 14220 can output the complex symbols through at least two paths. This can be modified by the designer.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions according to design.

The coding & modulation module according to another embodiment of the present invention, illustrated in FIG. 14, can output data pipes, PLS-pre information and PLS-post information processed for respective paths to the frame structure module.

FIG. 15 illustrates a demapping & decoding module according to another embodiment of the present invention.

The demapping & decoding module shown in FIG. 15 corresponds to another embodiment of the demapping & decoding module illustrated in FIG. 11. The demapping & decoding module shown in FIG. 15 can perform a reverse operation of the operation of the coding & modulation module illustrated in FIG. 14.

As shown in FIG. 15, the demapping & decoding module according to another embodiment of the present invention can include a first block 15000 for SISO, a second block 11100 for MISO, a third block 15200 for MIMO and a fourth block 14300 for processing the PLS-pre/PLS-post information. In addition, the demapping & decoding module can include blocks for processing data pipes equally or differently according to design. The first to fourth blocks 15000 to 15300 shown in FIG. 15 are similar to the first to fourth blocks 11000 to 11300 illustrated in FIG. 11.

However, the first to fourth blocks 15000 to 15300 shown in FIG. 15 are distinguished from the first to fourth blocks 11000 to 11300 illustrated in FIG. 11 in that an I/Q deinterleaver and derotation block 15010 is present between the time interleaver and the cell deinterleaver of the first to fourth blocks 15000 to 15300, a constellation mapper 15010 included in the first to fourth blocks 15000 to 15300 has a function different from the first to fourth blocks 11000 to 11300 illustrated in FIG. 11 and the third block 15200 for MIMO has a configuration different from the third block 11200 for MIMO illustrated in FIG. 11. The following description focuses on these differences between the first to fourth blocks 15000 to 15300 shown in FIG. 15 and the first to fourth blocks 11000 to 11300 illustrated in FIG. 11.

The I/Q deinterleaver & derotation block 15010 can perform a reverse process of the process performed by the rotation & I/Q interleaver block 14020 illustrated in FIG. 14. That is, the I/Q deinterleaver & derotation block 15010 can deinterleave I and Q components I/Q-interleaved and transmitted by the apparatus for transmitting broadcast signals and derotate complex symbols having the restored I and Q components.

The I/Q deinterleaver & derotation block 15010 is commonly applicable to the first to fourth blocks 15000 to 15300, as described above. In this case, whether or not the I/Q deinterleaver & derotation block 15010 is applied to the fourth block 15300 for processing the PLS-pre/post information can be signaled through the above-described preamble.

The constellation demapper block 15020 can perform a reverse process of the process performed by the constellation mapper block 14010 illustrated in FIG. 14. That is, the constellation demapper block 15020 can demap cell-deinterleaved data without performing derotation.

The third block 15200 for MIMO can include a complex symbol parsing block 15210 and a Q-block deinterleaver block 15220, as shown in FIG. 15.

The complex symbol parsing block 15210 can perform a reverse process of the process performed by the complex symbol generator block 14220 illustrated in FIG. 14. That is, the complex symbol parsing block 15210 can parse complex data symbols and demap the same to bit data. In this case, the complex symbol parsing block 15210 can receive complex data symbols through at least two paths.

The Q-block deinterleaver block 15220 can perform a reverse process of the process carried out by the Q-block interleaver block 14210 illustrated in FIG. 14. That is, the Q-block deinterleaver block 15220 can restore Q size blocks according to row-column deinterleaving, restore the order of permuted blocks to the original order and then restore positions of parity bits to original positions according to parity deinterleaving.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions according to design.

As illustrated in FIG. 15, the demapping & decoding module according to another embodiment of the present invention can output data pipes and PLS information processed for respective paths to the output processor.

As described above, the apparatus and method for transmitting broadcast signals according to an embodiment of the present invention can multiplex signals of different broadcast transmission/reception systems within the same RF channel and transmit the multiplexed signals and the apparatus and method for receiving broadcast signals according to an embodiment of the present invention can process the signals in response to the broadcast signal transmission operation. Accordingly, it is possible to provide a flexible broadcast transmission and reception system.

As described above, the waveform generation module 1300 according to an embodiment of the present invention may convert signal frames output from the frame structure module 1200 into ultimately transmittable signals. In this case, the waveform generation module 1300 according to an embodiment of the present invention may use a phase pre-distortion (PPD) method (or phase distortion). The phase pre-distortion method according to an embodiment of the present invention may be also referred to as a distributed MISO (Multi Input Single Output) scheme or 2D-eSFN. In addition, the present invention assumes that input signals of the waveform generation block 1300 are the same.

The system according to the present invention supports the SFN (Single Frequency Network) network, where distributed MISO scheme is optionally used to support very robust transmission mode. The 2D-eSFN is a distributed MISO scheme that uses multiple TX antennas, each of which is located in the different transmitter site in the SFN network. The each of the multiple TX (Transmitter) antennas is identified according to Tx ID.

In the SFN configuration, the 2D-eSFN processing independently distorts the phase of the signals transmitted from multiple transmitters, in order to create both time and frequency diversity. Hence, burst errors due to low flat fading or deep-fading for a long time can be mitigated.

According to the phase pre-distortion method of the present invention, the performance of channel estimation by a broadcast signal reception apparatus may not deteriorate and gain distortion of a transmission signal may not be caused and thus the loss of transmission capacity due to the gain distortion may be minimized.

In addition, the phase pre-distortion method of the present invention may be applied independently to a plurality of TX antennas as described above and thus a diversity gain may be achieved. Further, since the broadcast signal reception apparatus does not need to process phase pre-distortion, additional complexity is not required to design the broadcast signal reception apparatus.

In general, MISO processing is used to acquire diversity gain by using multiple antennas while causing less overhead and lower complexity.

The present invention proposes the 2D-eSFN as an example of the MISO scheme described above. Three schemes below of acquiring phase variation for diversity gain are proposed as the 2D-eSFN according to the example of the present invention.

A first scheme is an improved inter-symbol scheme, and is an improved scheme of performing phase transition within a symbol in the existing 2D-eSFN technology.

A second scheme is an intra-frame scheme, and is a scheme of performing phase transition by changing a phase sequence for each unit of a signal frame (or frame) using a particular pattern or a Tx-ID sequence of a frequency dimension (or domain).

A third scheme is an inter-symbol scheme, and is a scheme of performing phase transition by changing a phase in symbol units.

Each of the first scheme and the second scheme described above is a scheme for increasing an eSFN in the frequency dimension, and the third scheme is a scheme for performing phase transition through additional phase variation in the time dimension.

Hereinafter, effects of the first scheme and the second scheme in the frequency dimension are briefly described.

The first scheme described above may include change of the method of the phase transition between subcarriers (improved intra-symbol phase variation). When the first scheme is employed, it is possible to acquire better flatness with respect to the magnitude of the spectrum after eSFN processing.

The second scheme described above may include frame-wise cyclic Tx-ID sequence shift (left/right) (inter-frame phase variation). When the second scheme is employed, it is possible to acquire additional phase variation in the frequency dimension.

Hereinafter, an effect of the third scheme in the time dimension is briefly described.

The third scheme described above may include symbol-wise phase variation (inter-symbol phase variation). When the third scheme is employed, it is possible to acquire an advantage in the slow fading nature. Details are described below.

FIG. 16 is a view illustrating a waveform generation module according to another embodiment of the present invention.

The waveform generation module illustrated in FIG. 16 corresponds to another embodiment of the waveform generation module 1300 described above in relation to FIGS. 1 and 7.

As described above, the waveform generation module according to an embodiment of the present invention may receive signal frames output from the frame structure module described above in relation to FIG. 6 and modulate the received signal frames to correspond to the number of antennas to output the modulated signal frames.

That is, the waveform generation module corresponds to an embodiment of a waveform generation module a transmission apparatus using m Tx antennas, and may include m processing blocks for modulating input frames by m paths and outputting the modulated frames. The m processing blocks may perform the same processing procedure.

Each processing block included in the waveform generation module illustrated in FIG. 16 may include a reference signal insertion & PAPR reduction block, a FD (Frequency Dimension) phase pre-distortion block 16000, pattern generation & control block 16010, an inverse waveform transform block, a PAPR reduction in time block, a guard sequence insertion block, a preamble insertion block, a TD (Time Dimension) phase pre-distortion block 16020, a waveform processing block, an other system insertion block and a Digital Analog Conveter (DAC) block. The processing blocks included in the waveform generation module illustrated in FIG. 16 are the same as the processing blocks included in the waveform generation module illustrated in FIG. 7 except that the FD phase pre-distortion block 16000, the pattern generation & control block 16010 and the TD phase pre-distortion block 16020 in FIG. 16.

Accordingly, operations and functions of the blocks other than the FD phase pre-distortion block 16000, the pattern generation & control block 16010 and the TD phase pre-distortion block 16020 are the same as those described above in relation to FIG. 7 and thus are not described here, and a description is now given of t the FD phase pre-distortion block 16000, the pattern generation & control block 16010 and the TD phase pre-distortion block 16020 only.

The FD phase pre-distortion block 16000 may perform the first scheme and the second scheme described above corresponding to phase transition between phase values generated from a Tx-ID or a particular pattern. The phase generation & control block 16010 may generate a particular pattern or a Tx-ID for phase variation, and may change the generated pattern or Tx-ID for each unit of a symbol or a signal frame. The TD phase pre-distortion block 16020 may perform the above-described third scheme of performing phase transition through additional phase variation in symbol units in the time dimension. The TD phase pre-distortion block 16020 may perform phase change or phase hold-up using the Tx-ID or the particular pattern generated by the phase generation & control block 16010 described above.

Therefore, the broadcast signal transmission apparatus according to embodiments of the present invention may enhance a reception rate of the broadcast signal reception apparatus by performing phase pre-distortion differently between antennas before transmitting a signal to a Tx antenna. It is possible to obtain a high reception rate as a result of transmit antenna diversity gain due to the phase pre-distortion according to embodiments of the present invention.

Positions, terms, functions, etc. of the FD phase pre-distortion block 16000, the phase generation & control block 16010, and the TD phase pre-distortion block 16020 may be changed according to intention of a designer.

FIG. 17 is graphs related to the phase transition according to the first scheme of the present invention.

FIG. 17( a) is a graph illustrating a 2D-plot of an FD filter according to the existing 2D-eSFN technology, and FIG. 17( b) is a graph illustrating a 2D-plot of an FD filter of the first scheme according to embodiments of the present invention.

The existing 2D-eSFN technology may divide orthogonal frequency division multiplexing (OFDM) subcarriers into certain intervals, and determine a phase to be applied to each set Tx-ID. A phase for each interval may be set to a unique phase multiplied by a raised cosine function for an interval.

The graphs at the top of FIGS. 17( a) and (b) illustrate spectrums in frequency dimensions corresponding to results of phase distortions according to the existing 2D-eSFN technology and the first scheme according to embodiments of the present invention, respectively. A horizontal axis of the graphs indicates subcarrier indices of OFDM, and a vertical axis thereof indicates powers and magnitudes. A spectrum of the frequency dimension corresponds to a unique phase multiplied by a raised cosine function described above.

According to the existing 2D-eSFN technology as described in FIG. 17( a), ripples may be generated in the frequency-dimensional spectrum. However, according to the first scheme of the present invention as described in FIG. 17( b), ripples may disappear from the frequency-dimensional spectrum.

When the first scheme according to embodiments of the present invention is employed, direct phase transition may be performed between at least two phases changed for each interval. Therefore, the first scheme according to embodiments of the present invention may acquire better flatness with respect to the magnitude of the spectrum after eSFN processing when compared to the existing 2D-eSFN technology.

Two graphs in the middle of each of FIGS. 17( a) and (b) are graphs depicting raised cosine functions, to which phase sequences are applied, as real values and imaginary values, respectively for subcarriers of OFDM.

A graph at the bottom of each of FIGS. 17( a) and (b) is a graph illustrating the graph at the top thereof in 2-D coordinates. The graph shows that phase transition occurs along a circular arc having a magnitude of 1 in each of the above-described schemes. Specific description of the first scheme is given below.

FIG. 18 is a conceptual diagram of phase transition according to the second scheme of the present invention.

As described above, the second scheme of the present invention is a scheme of performing phase transition by cyclic-shifting a Tx-ID sequence applied for each Tx in a unit of a signal frame. The Tx-ID sequence according to embodiments of the present invention may be shifted in the left direction or the right direction at a frame level.

FIG. 18 illustrates operations of calculating phase sequence indices from signal frame 0 to signal frame n when the length of a TX-ID sequence is 8, and a cyclic shift direction is different between Tx1 and Tx2. Specifically, FIG. 18( a) illustrates an operation of calculating a phase sequence index for each frame when Tx1 is cyclic-shifted in the left direction, and FIG. 18( b) illustrates an operation of calculating a phase sequence index for each frame when Tx2 is cyclic-shifted in the right direction.

As illustrated in FIG. 18, when a Tx-ID sequence is cyclic-shifted in different directions, a relative phase of the sequence may change between respective Txs in each signal frame. Therefore, it is possible to acquire additional phase variation of the frequency dimension.

FIG. 19 illustrates a phase transition operation according to the third scheme of the present invention.

As described in the foregoing, the third scheme is a scheme of performing phase transition by adding a phase variation to a basic phase in symbol units in the time dimension. According to the third scheme of the present invention, it is possible to prevent broadcast signals from mutually attenuating for a long period of time by using different symbol-based phase variation patterns for each Tx in slow fading.

FIG. 19 illustrates phase variation of each symbol when a phase is changed in symbol units within a signal frame (for example, Nth signal frame). Referring to FIG. 19, the horizontal axis indicates each symbol, and the vertical axis indicates a changed phase value corresponding to each symbol.

Phase values may have a minimum value of (1) p_base, and may have a maximum value of (2) p_base+p_var. The value p_base indicates a basic phase, and the value p_var indicates a phase value changed from the basic phase.

Referring to FIG. 19, (3) phase for the symbol-wise phase variation corresponds to a value sampled from a sine wave, and may have the same value within a symbol. An example of phase variation of the present invention may use only a sine wave. When a Tx-ID sequence or another particular phase pattern is used in addition to a sine wave, the broadcast signal transmission apparatus according to embodiments of the present invention may perform hold-up or transition of a phase using (4) section of sine wave, that is, by dividing a sine wave into a plurality of sections as described in FIG. 19. In this case, the phase transition may be performed using sections corresponding to values from a maximum value to a minimum value or from a minimum value to a maximum value of a sine wave. Therefore, as illustrated in FIG. 19, phase transition may be performed in a first section, and phase hold-up may be performed in a second section.

In addition, to obtain continuity of channel estimation between signal frames, the broadcast signal transmission apparatus according to embodiments of the present invention may set an initial symbol and a last symbol of a signal frame to have the same phase value as illustrated in FIG. 19, or may sample continuous values from the sine wave. The setting may be changed according to intention of a designer.

FIG. 20 is a conceptual diagram of phase transition in the frequency dimension according to a conventional technology.

FIG. 20( a) is coordinates indicating phases, and FIG. 20( b) indicates a section to which each phase is applied and a phase value corresponding to each section. FIG. 20( c) is a graph indicating a real value (cos(A), cos(B)) of a phase corresponding to each section, and a graph indicating an imaginary value (sin(A), sin(B)) of a phase corresponding to each section. FIG. 20( d) is a graph indicating power variation of a transmitted signal due to phase pre-distortion.

When a phase pattern is changed according to embodiments of the present invention, a corresponding phase value may change accordingly. FIG. 20( a) illustrates a position of each of phase A and phase B in a real coordinate and an imaginary coordinate when phases according to embodiments of the present invention include phase A and phase B. In this case, coordinates of phase A may be expressed by (cos(A), sin(A)), and coordinates of phase B may be expressed by (cos(B), sin(B)). When a phase is directly changed from phase A to phase B or from phase B to phase A, the broadcast signal transmission apparatus according to embodiments of the present invention may perform phase transition or linear transition along a straight line corresponding to a shortest distance between cos(A) and cos(B), or a straight line corresponding to a shortest distance between sin(A) and sin(B).

A section illustrated in FIG. 20( b) is obtained by dividing OFDM subcarriers into certain sections, may be referred to as an FD sub-block, and is a unit section for performing of phase transition. As described in the foregoing, a phase of each section may be determined according to a Tx-ID set for each section. In particular, a phase for each section of an eSFN may be obtained by multiplying a phase by a raised cosine function for the section.

FIG. 20( c) illustrates real and imaginary values of a raised cosine function to which a phase is applied for each section in a unit of a subcarrier. As illustrated in FIG. 20( c), real values (cos(A), cos(B)) and imaginary values (sin(A), sin(B)) of each phase may have values between a minimum value 0 and a maximum value 1. In addition, when a value of each phase is changed at a boundary of each section, a real value and an imaginary value of the phase are immediately changed to a real value and an imaginary value of a phase corresponding to a subsequent section.

As described with reference to FIG. 20( a), when a phase is directly changed from phase A to phase B or from phase B to phase A, the broadcast signal transmission apparatus according to embodiments of the present invention may perform phase transition or linear transition along a straight line corresponding to a shortest distance between cos(A) and cos(B), or a straight line corresponding to a shortest distance between sin(A) and sin(B). In an example of the present invention, a section in which phase distortion is performed when a phase is changed from phase A to phase B is referred to as PTP 1, and a section in which phase distortion is performed when a phase is changed from phase B to phase A is referred to as PTP 2. As illustrated in FIG. 20, when phase transition is performed along a straight line, power of a transmitted signal is less than 1 in sections PTP1 and PTP2. That is, ripples are generated in the frequency dimension. Therefore, a reception rate of the broadcast signal reception apparatus is degraded.

FIG. 21 is a conceptual diagram illustrating phase transition according to embodiments of the present invention.

The phase transition according to embodiments of the present invention may be directly performed using linear interpolation between two phase values instead of phase transition using a raised cosine function described with reference to FIG. 20.

When a phase is directly changed from phase A to phase B or from phase B to phase A, the phase transition according to embodiments of the present invention may be performed along a curve (2) connecting (cos(A), sin(A)) to (cos(B), sin(B)) instead of a straight line (1) corresponding to a shortest distance between cos(A) and cos(B) or corresponding to a shortest distance between sin(A) and sin(B). In this case, power of a transmitted signal may be maintained at 1 in a section in which a phase is changed.

In the phase transition according to embodiments of the present invention, it is possible to deduce Dout(k) by multiplying an input signal Din(k) by a complex signal e^(jθ(k)) using a phase that undergoes linear interpolation in the frequency dimension. The phase transition according to embodiments of the present invention may provide better flatness with respect to the magnitude of the spectrum compared with the eSFN and reduce the relative power of the side lobe in the time-domain CIR (channel impulse response) of the FD raised cosine function.

FIG. 22 illustrates phase transition according to the first scheme of the present invention.

FIG. 22( a) illustrates a section to which each phase is applied and a phase value corresponding to each section. FIG. 22( b) is a graph illustrating real values (cos(A), cos(B)) of phases corresponding to respective sections, and a graph illustrating imaginary values (sin(A), sin(B)) of phases corresponding to respective sections. FIG. 22( c) is a graph illustrating power variation of a transmitted signal due to phase pre-distortion according to the first scheme of the present invention.

Description of FIG. 22( a) is similar to the description of FIG. 20( b), and is different from the description of FIG. 20( b) in the following point.

According to the first scheme of the present invention, starting points of the sections PTP1 and PTP2 are defined as k1s and k2s, respectively, and ending points of the respective sections are defined as k1e and k2e, respectively. Hereinafter, as described above, the broadcast signal transmission apparatus according to embodiments of the present invention may acquire phase θ(k) in a transition interval between phase A and phase B based on a (3) linear phase interpolation equation. Thereafter, the broadcast signal transmission apparatus according to embodiments of the present invention may convert a phase value acquired with respect to a previous subcarrier into a complex signal for phase rotation.

As described in FIG. 22( b), when a phase is directly changed from phase A to phase B, the broadcast signal transmission apparatus according to embodiments of the present invention may perform phase distortion along a curve connecting cos(A) to cos (B) or a curve connecting sin(A) to sin(B) for an interval of PTP1. In this case, the broadcast signal transmission apparatus according to embodiments of the present invention may acquire Dout(k) by multiplying the converted complex signal by Din(k). Similar description is applied to an interval of PTP2.

Accordingly, as illustrated in FIG. 22(C), when phase pre-distortion is performed according to the first scheme of the present invention, power of a transmitted signal may be maintained at 1 in the intervals PTP1 and PTP2.

FIG. 23 illustrates another example the conceptual diagram of phase transition according to the second scheme of the present invention.

FIG. 23 illustrates operations of calculating phase sequence indices from signal frame 0 to signal frame n when the length of a TX-ID sequence is L, and a cyclic shift direction is different between Tx1 and Tx2. The shift direction may be different between adjacent transmitters. A unit of the shift may be set to various values, and a unit of the shift may be different between Txs. The unit may be changed according to intention of a designer.

FIG. 23( a) illustrates an operation of calculating a phase sequence index for each frame when Tx1 is cyclic-shifted in the left direction, and FIG. 23( b) illustrates an operation of calculating a phase sequence index for each frame when Tx2 is cyclic-shifted in the right direction.

An equation shown at the bottom of each of FIGS. 23( a) and 23(b) indicates a shifted Tx-ID sequence of a signal frame indicated by a frame index for each shift direction. An equation indicating a Tx-ID sequence index indicates a shifted Tx-ID sequence at an nth frame from Tx1 and Tx2 each corresponding to an original Tx-ID sequence. In addition, a cyclic shift according to embodiments of the present invention may be reset for each initial signal frame of a super frame.

FIG. 24 illustrates another example of the phase transition operation according to the third scheme of the present invention.

FIG. 24 is another example of the third scheme described in FIG. 19. FIG. 24 is similar to FIG. 19, and is different from FIG. 19 in that (3) phase for the symbol-wise phase variation is indicated by p(s), and (5) hold-up period and (6) segment are indicated. In addition, FIG. 24( b) shows equations and parameters indicating the phase transition according to the third scheme of the present invention.

As described above, the third scheme of the present invention may be divided into a case of performing phase transition using a sine wave, and a case of performing phase transition using both a sine wave and a Tx-ID sequence.

The equations shown on the right side of FIG. 24( b) indicate the third scheme using only a sine wave described above. The equations shown in FIG. 24( b) indicate operations of calculating p(s), a phase for the symbol-wise phase variation, an output signal, and Dout(s) using values corresponding to the parameters shown on the left side of FIG. 24( b). Din(s) and Dout(s) are signal sequences of an sth symbol, and Dout(s) corresponds to a value obtained by multiplying a complex signal e^(jθ(k)) by an input signal Din(k). The length of each signal sequence is the same as the length of an OFDM symbol, and the length of each OFDM symbol may include the length of a guard interval.

As described in the foregoing, in the third scheme using both a sine wave and a Tx-ID sequence, a phase hold-up function may be used while a TX-ID sequence or a random sequence is used. In this case, a start phase (p_start) may be different with respect to the transmitter. In addition, a (5) hold-up period illustrated in FIG. 24( a) may be generated depending on a Tx-ID sequence used. When a Tx-ID sequence is used, the broadcast signal transmission apparatus according to embodiments of the present invention may determine a symbol section (segment) controlled by the Tx-ID sequence. The length of the segment may be set to an arbitrary constant length, and may be expressed by (6) segment as illustrated in FIG. 24( a). The segment of the present invention may be divided into a hold-up segment and a phase up/down segment according to a Tx-ID sequence used.

The phase up segment according to embodiments of the present invention may be used after adding 1.0 to a value in a range of −cos(0°) to −cos(180°), and performing normalization, and the phase down segment may be used after adding 1.0 to a value in a range of cos(0°) to cos(180°), and performing normalization. When the Tx-ID sequence is used, and the number of segments is greater than the length of the Tx-ID sequence, a Tx-ID may be repeatedly used.

When R (corresponding to the number of the repetition of the sine wave in a frame) is an integer, the phase can change seamlessly at the frame edge.

FIG. 25 illustrates a waveform generation module according to another example of the present invention.

FIG. 25 corresponds to another example of the waveform generation module described in FIGS. 1, 7, and 16.

As described in the foregoing, the waveform generation module according to embodiments of the present invention may receive inputs of signal frames output from the frame structure module described in FIG. 6, and modulate and transmit signal frames equal in number to the number of antennas for output.

The waveform generation module is an example of a waveform generation module of a transmitter using m Tx antennas, and may include m processing blocks for modulating and outputting input frames corresponding to m paths. The m processing blocks may perform the same processing operation.

The waveform generation module shown in FIG. 25 includes the same blocks as those of FIG. 16, and is different from the waveform generation module of FIG. 16 in that a TD phase pre-distortion block is not positioned after an inverse waveform transform block, and an FD phase pre-distortion block performs TD phase pre-distortion. That is, phase pre-distortions (or phase transitions) in the frequency and time dimensions are performed before inverse waveform transformation. Hereinafter, phase pre-distortion in the frequency or time dimension performed before inverse waveform transformation may be referred to as 2D phase pre-distortion.

The 2D phase pre-distortion according to embodiments of the present invention may include all the above-described first to third schemes. However, another example may be applied to the third scheme corresponding to phase distortion in the time dimension. Details are described below.

FIG. 26 is an equation indicating 2D phase pre-distortion according to embodiments of the present invention.

FIG. 26( a) is an equation indicating an input signal and an output signal of 2D phase pre-distortion according to embodiments of the present invention.

FIG. 26( b) is an equation for obtaining a frequency-dimensional phase value of 2D phase pre-distortion according to embodiments of the present invention.

FIG. 26( c) is an equation for obtaining a time-dimensional phase value of 2D phase pre-distortion according to embodiments of the present invention.

The equation shown in FIG. 26( a) includes both phase values in the time dimension and the frequency dimension. In the SFN configuration, the 2D-eSFN processing independently distorts the phase of the signals transmitted from multiple transmitters, in order to create both time and frequency diversity. Hence, burst errors due to low flat fading or deep-fading for a long time can be mitigated. The time-frequency dimensional phase-distorted signals are as shown in (a) and where q_(m,l,k) and r_(m,l,k) are the input and output signals of 2D-eSFN processing, respectively. The phase-distortion value for the k^(th) carrier of the l^(th) symbol and is defined as shown in (a). The phase-distortion value is calculated based on the time- and frequency-dimensional phase terms, respectively.

When TD phase pre-distortion is performed after inverse waveform transformation as described in FIG. 16, θ_(t)(l) corresponding to a time-dimensional phase value may be excluded from the equation shown in FIG. 26( a), and exp(θ_(t)(l)) may be multiplied after preamble insertion.

The frequency-dimensional phase term for the k^(th) carrier is calculated as shown in (b) and the time-dimensional phase term for the l^(th) symbol carrier is calculated as shown in (c).

A portion corresponding to (A) in the equation shown in FIG. 26(C) may be replaced by another expression having an arbitrary waveform. The replacement may be changed according to intention of a designer.

FIG. 27 illustrates a phase transition operation according to the third scheme of the 2D phase pre-distortion according to embodiments of the present invention.

FIG. 27 is another example of the third scheme described in FIGS. 19 and 24, and indicates a case applied to the 2D phase pre-distortion according to embodiments of the present invention. As described in the foregoing, the 2D phase pre-distortion according to embodiments of the present invention is different from the third scheme described with reference to FIGS. 19 and 24 since phase distortion in the time dimension is performed before inverse waveform transformation.

Specifically, FIG. 27( a) is similar to FIG. 24( a), and is different from FIG. 24( a) in that a hold-up period is absent.

FIG. 27( b) shows an equation and parameters indicating phase transition according to the third scheme of the 2D phase pre-distortion according to embodiments of the present invention. The equation shown in FIG. 27( b) is an example of using a sinusoidal wave for phase variation. However, it is possible to use an arbitrary wave, for example, a saw tooth, a saw tooth or a sinusoidal wave having a variable periodicity, etc. The wave may be changed according to intention of a designer.

FIG. 28 is a flowchart illustrating a method for transmitting broadcast signals according to an embodiment of the present invention.

The apparatus for transmitting broadcast signals according to an embodiment of the present invention can encode service data (S28000). As described above, service data is transmitted through a data pipe which is a logical channel in the physical layer that carries service data or related metadata, which may carry one or multiple service(s) or service component(s). In the present invention, the data pipe can be referred as a transmission unit for robustness control, thereby affecting quality-of-service (QoS). Data carried on a data pipe can be referred to as the DP data or the service data. The detailed process of step S24000 is as described in FIG. 1 or 5 or 14.

The apparatus for transmitting broadcast signals according to an embodiment of the present invention can build at least one signal frame including the encoded service data (S28010). The detailed process of this step is as described in FIG. 6.

Then, the apparatus for transmitting broadcast signals according to an embodiment of the present invention can perform a phase distortion on data in the built at least one signal frame (S28020). As above described, in the SFN configuration, the 2D-eSFN processing independently distorts the phase of the signals transmitted from multiple transmitters, in order to create both time and frequency diversity. Hence, burst errors due to low flat fading or deep-fading for a long time can be mitigated. The apparatus for transmitting broadcast signals according to an embodiment of the present invention may perform the phase distortion data in both time and frequency dimension. The detailed process of the phase distortion is as described in FIGS. 16 to 27.

Subsequently, the apparatus for transmitting broadcast signals according to an embodiment of the present invention can modulate data in the built at least one signal frame by an OFDM (Orthogonal Frequency Division Multiplexing) scheme (S28030). The detailed process of this step is as described in FIG. 1 or 7.

The apparatus for transmitting broadcast signals according to an embodiment of the present invention can transmit the broadcast signals including the modulated data (S28040). The detailed process of this step is as described in FIG. 1 or 7.

FIG. 29 is a flowchart illustrating a method for receiving broadcast signals according to an embodiment of the present invention.

The flowchart shown in FIG. 29 corresponds to a reverse process of the broadcast signal transmission method according to an embodiment of the present invention, described with reference to FIG. 28.

The apparatus for receiving broadcast signals according to an embodiment of the present invention can receive broadcast signals (S29000). A phase of the broadcast signals according to an embodiment of the present invention may be distorted in a transmission side. As above described, in the SFN configuration, the 2D-eSFN processing independently distorts the phase of the signals transmitted from multiple transmitters, in order to create both time and frequency diversity. Hence, burst errors due to low flat fading or deep-fading for a long time can be mitigated. The apparatus for transmitting broadcast signals according to an embodiment of the present invention may perform the phase distortion data in both time and frequency dimension. The detailed process of the phase distortion is as described in FIGS. 16 to 27.

The apparatus for receiving broadcast signals according to an embodiment of the present invention can demodulate the received broadcast signals using an OFDM (Othogonal Frequency Division Multiplexing) scheme (S29010). Details are as described in FIG. 8 or 9.

The apparatus for receiving broadcast signals according to an embodiment of the present invention can parse at least one signal frame from the demodulated broadcast signals (S29020). Details are as described in FIG. 8 or 10.

Subsequently, the apparatus for receiving broadcast signals according to an embodiment of the present invention can decode data in the parsed at least one signal frame to output service data (S29030). Details are as described in FIG. 8 or 11 and FIG. 16. As described above, service data is transmitted through a data pipe which is a logical channel in the physical layer that carries service data or related metadata, which may carry one or multiple service(s) or service component(s). In the present invention, the data pipe can be referred as a transmission unit for robustness control, thereby affecting quality-of-service (QoS). Data carried on a data pipe can be referred to as the DP data or the service data.

It will be apparent to those skilled in the art that various modifications and variations can be made in the present invention without departing from the spirit or scope of the invention. Thus, it is intended that the present invention covers the modifications and variations of this invention provided they come within the scope of the appended claims and their equivalents. 

What is claimed is:
 1. A method for transmitting broadcast signals, the method comprising: encoding service data corresponding to each of a plurality of transmission unit, wherein the each of a plurality of transmission unit carries at least one service component; building at least one signal frame including the encoded service data; performing a phase distortion on data in the built at least one signal frame; modulating data in the built at least one signal frame by an OFDM (Orthogonal Frequency Division Multiplex) scheme; and transmitting the broadcast signals including the modulated data.
 2. The method of claim 1, wherein the phase distortion is performed in a time dimension and a frequency dimension.
 3. The method of claim 2, wherein the phase distortion is performed according to a phase distortion value, wherein the phase distortion value comprises a time dimensional phase term and a frequency dimensional phase term.
 4. The method of claim 3, wherein the time dimensional phase term is calculated based on a base phase and a phase variation value for each symbol carrier.
 5. The method of claim 4, wherein the time dimensional phase term is expressed as: ${{\theta_{t}(l)} = {\theta_{base} + {\frac{1}{2}{\theta_{var}\left( {1 - {\cos \left( {\frac{2\pi \; l}{N_{R}} + ɛ_{offset}} \right)}} \right)}}}},$ wherein θ_(base)ε[0,2π) is the base phase, wherein θ_(var)ε[0,2π) is the phase variation value and wherein ε_(offset)ε[0,2π) is a phase offset for different apparatus for transmitting broadcast signals.
 6. An apparatus for transmitting broadcast signals, the apparatus comprising: an encoder for encoding service data corresponding to each of a plurality of transmission unit, wherein the each of a plurality of transmission unit carries at least one service component; a frame builder for building at least one signal frame including the encoded service data; a phase distortion unit for performing a phase distortion on data in the built at least one signal frame; a modulator for modulating data in the built at least one signal frame by an OFDM (Orthogonal Frequency Division Multiplex) scheme; and a transmitter for transmitting the broadcast signals including the modulated data.
 7. The apparatus of claim 6, wherein the phase distortion is performed in a time dimension and a frequency dimension.
 8. The apparatus of claim 7, wherein the phase distortion is performed based on a phase distortion value, wherein the phase distortion value comprises a time dimensional phase term and a frequency dimensional phase term.
 9. The apparatus of claim 8, wherein the time dimensional phase term is calculated based on a base phase and a phase variation value for each symbol carrier.
 10. The apparatus of claim 9, wherein the time dimensional phase term is expressed as: ${{\theta_{t}(l)} = {\theta_{base} + {\frac{1}{2}{\theta_{var}\left( {1 - {\cos \left( {\frac{2\pi \; l}{N_{R}} + ɛ_{offset}} \right)}} \right)}}}},$ wherein θ_(base)ε[0,2π) is the base phase, wherein θ_(var)ε[0,2π) is the phase variation value and wherein ε_(offset)ε[0,2π) is a phase offset for different apparatus for transmitting broadcast signals.
 11. A method for receiving broadcast signals, the method comprising: receiving the broadcast signals, wherein a phase of the broadcast signals is distorted; demodulating the received broadcast signals by an OFDM (Orthogonal Frequency Division Multiplex) scheme; parsing at least one signal frame from the demodulated broadcast signals; and decoding data in the parsed at least one signal frame to output service data, wherein the service data corresponds to each of a plurality of transmission unit in the parsed at least one signal frame, wherein the each of a plurality of transmission unit carries at least one service component.
 12. The method of claim 11, wherein the phase distortion is performed in a time dimension and a frequency dimension.
 13. The method of claim 12, wherein the phase distortion is performed according to a phase distortion value, wherein the phase distortion value comprises a time dimensional phase term and a frequency dimensional phase term.
 14. The method of claim 13, wherein the time dimensional phase term is calculated based on a base phase and a phase variation value for each symbol carrier.
 15. The method of claim 14, wherein the time dimensional phase term is expressed as: ${{\theta_{t}(l)} = {\theta_{base} + {\frac{1}{2}{\theta_{var}\left( {1 - {\cos \left( {\frac{2\pi \; l}{N_{R}} + ɛ_{offset}} \right)}} \right)}}}},$ wherein θ_(base)ε[0,2π) is the base phase, wherein θ_(var)ε[0,2π) is the phase variation value and wherein ε_(offset)ε[0,2π) is a phase offset for different apparatus for transmitting broadcast signals.
 16. An apparatus for receiving broadcast signals, the apparatus comprising: a receiver for receiving the broadcast signals, wherein a phase of the broadcast signals is distorted; a demodulator for demodulating the received broadcast signals by an OFDM (Orthogonal Frequency Division Multiplex) scheme; a frame parser for parsing at least one signal frame from the demodulated broadcast signals; and a decoder for decoding data in the parsed at least one signal frame to output service data, wherein the service data corresponds to each of a plurality of transmission unit in the parsed at least one signal frame, wherein the each of a plurality of transmission unit carries at least one service component.
 17. The apparatus of claim 16, wherein the phase distortion is performed in a time dimension and a frequency dimension.
 18. The apparatus of claim 17, wherein the phase distortion is performed according to a phase distortion value, wherein the phase distortion value comprises a time dimensional phase term and a frequency dimensional phase term.
 19. The apparatus of claim 18, wherein the time dimensional phase term is calculated based on a base phase and a phase variation value for each symbol carrier.
 20. The apparatus of claim 19, wherein the time dimensional phase term is expressed as: ${{\theta_{t}(l)} = {\theta_{base} + {\frac{1}{2}{\theta_{var}\left( {1 - {\cos \left( {\frac{2\pi \; l}{N_{R}} + ɛ_{offset}} \right)}} \right)}}}},$ wherein θ_(base)ε[0,2π) is the base phase, wherein θ_(var)ε[0,2π) is the phase variation value and wherein ε_(offset)ε[0,2π) is a phase offset for different apparatus for transmitting broadcast signals. 